Magnetic disk storage apparatus

ABSTRACT

Currents of sine waveforms can be fed through coils by a relatively small-sized circuit, and thereby, highly dense magnetic storage can be realized with less rotation variations and a driving control circuit of a motor rotating at a low noise level can be provided. A coil of one phase of a three-phase brushless motor is driven with full amplitude at which an applied voltage becomes equal to a source voltage, and a coil of one of other phases is driven with gradually changing voltages so that a current of sine waveform is delivered, and a coil of the remaining phase is driven by feedback control so that a total current flowing through all coils becomes a predetermined current value.

BACKGROUND OF THE INVENTION

[0001] The present invention relates to technology for driving controlof a brushless motor, and more particularly to technology effectivelyapplied to the formation of rotation drive current waveforms of themotor. The present invention relates to technology effectively appliedto a driving control apparatus of a spindle motor for rotationallydriving disk type storage media as in, e.g., a hard disk drive.

[0002] A hard disk drive is demanded to have the ability to read andwrite information from and to magnetic disk as fast as possible, thatis, the ability to make access at high speed. To achieve this, it isimportant to speed up disk rotation. Conventionally, a brushless DCmulti-phase motor called a spindle motor has been generally used torotate magnetic disk in a hard disk drive. The magnetic disk is fastrotated by the spindle motor and a magnetic head for read and writing isbrought near to the surface of the rotating magnetic disk to write orread information while moving in a radius direction thereof.

[0003] In rotation driving control of a conventional spindle motor, arotor has been rotated by supplying coils of individual phases withsquare-wave pulse currents as shown in FIG. 15 that are out of phasewith one another, by a driving circuit. FIG. 15 shows the waveform ofcurrent fed through one of three phases; currents having waveforms thatare 120 degrees out of phase with one another are fed through other twophases. Such a rotation driving method based on square-wave pulsecurrents has the advantage of easy current formation but also thedisadvantage of causing rotation variations and noise due to torqueripple. It is known that a brushless motor can be rotated withoutcausing rotation variations and noise by using drive current waveformsof sine waveforms. Accordingly, an invention is proposed which smoothlyrotates a rotor by feeding pulse currents of sine waveforms throughcoils of individual phases (Japanese Published Unexamined PatentApplication No. Hei 9(1997)-37584).

SUMMARY OF THE INVENTION

[0004] However, in the above described technology, plural units ofwaveform information of one cycle of current waveforms to be formed arestored in ROM (read only memory), depending on the load on the motor,and when a user selectively specifies one of them, the specifiedwaveform information is read out to control coil drive currents, wherebycurrents of desired sine waveforms are outputted. As a result, theamount of hardware increases, and even if the load on the motor changes,since the duty of basic clock to form coil drive waveforms remainsconstant, phase switching of output currents cannot be smoothlyperformed in response to an increase or decrease in the output currents.This fact has been revealed by the present inventors.

[0005] An object of the present invention is to provide a magnetic diskunit that can feed currents of sine waveforms through coils by arelatively small-sized circuit, and thereby, enables highly densemagnetic storage with less rotation variations and has a spindle motorrotating at a low noise level.

[0006] Another object of the present invention is to provide a magneticdisk unit that can smoothly change output currents in response tochanges in the load on a motor, and thereby, enables highly densemagnetic storage with less rotation variations and has a spindle motorrotating at a low noise level.

[0007] The above described objects and other objects and characteristicsof the present invention will become apparent from the description ofthis specification and the accompanying drawings.

[0008] Typical ones of intentions disclosed by the present patentapplication will be briefly described below.

[0009] A magnetic disk storage apparatus of this invention comprises: afirst motor for rotating magnetic disk; a magnetic head for readinginformation from recording tracks on the magnetic disk; and a firstmotor driving control circuit for controlling drive currents of thefirst motor, wherein the first motor is a multi-phase brushless motor inwhich the potential of a center tap of the multi-phase brushless motoris made to be floating, and a driving control circuit of the first motorperforms driving by feedback control so that a coil of one of the phasesis driven with a full amplitude at which an applied voltage becomesequal to a source voltage, a coil of a second phase is driven withgradually changing voltages so that a current of sine waveform isdelivered, and a third coil is controlled so that a total currentflowing through all coils becomes a predetermined current value.

[0010] According to the above described means, motor coils can be drivenaccording to sine waveforms without causing power loss, whereby diskrotation variations are reduced, highly dense magnetic storage isenabled, and the motor can rotate at a low noise level.

[0011] Preferably, the first motor driving control circuit is providedwith an arithmetic circuit that produces by predetermined operations asignal driven with gradually changing voltages so that a current of sinewaveform is delivered. Accordingly, in comparison with the method ofholding all data corresponding to sine waveforms in memory, a circuitscale can be made smaller and the magnetic disk storage apparatus can beminiaturized.

[0012] Moreover, the first motor driving control circuit is constructedto produce as a PWM signal a signal driven with gradually changingvoltages so that a current of sine waveform is delivered. A drivingmethod based on the PWM signal enables less power loss than a drivingmethod based on linearly changing currents.

[0013] The first motor driving control circuit is constructed to produceas a PWM signal a signal driven with the feedback control. Use of thePWM signal can reduce power loss and enables still less rotationvariations because it can be driven with currents corresponding tochanging loads.

[0014] Moreover, coil currents fed through coils of individual phases bythe first motor driving control circuit are formed to have phases thatare an predetermined electrical angle corresponding to coil inductanceand internal resistance ahead of the phases of back electromotive forcesinduced in the coils. Accordingly, the motor can be rotated with thegreatest driving torque.

[0015] Moreover, the first motor driving control circuit drives coils ofindividual phases so that phase switching timing is off zero-crosspoints of the back electromotive forces. Thereby, in the case wherephase switching control is performed by detecting zero-cross points ofback electromotive forces, the detection of incorrect zero-cross pointsdue to noise generated in the coils during phase switching can beprevented, so that highly accurate rotation control can be performed.

[0016] The first motor driving control circuit produces signals drivenwith gradually changing voltages by identical operations even if phasesdriven by the signals are different from each other so that currents ofsine waveforms are delivered. By producing drive control signals of allphases by identical operations, circuit configuration and arithmeticprograms can be simplified.

[0017] Moreover, in a magnetic disk storage apparatus comprising thefirst motor driving control circuit and a controller controlling thefirst motor driving control circuit, the first motor driving controlcircuit is constructed to perform control so that the total of currentsfed through the coils of the phases matches a current indication valuesupplied from the controller, and a current indication value correctingcircuit is provided which corrects the current indication value, takinginto account fluctuations of the total current produced by the currentsfed through the coils of the phases being changed according to sinewaveforms. Accordingly, reaction of the control system to ripples ofcoil current resulting from driving the motor with a sine waveform canbe weakened, with the result that torque ripples can be reduced androtation variations can be further lessened.

BRIEF DESCRIPTION OF THE DRAWINGS

[0018]FIG. 1 shows a driving circuit in a three-phase brushless motor towhich the present invention is effectively applied, and an equivalentcircuit of the motor;

[0019]FIG. 2 illustrates vector representation of applied voltageVinput, coil voltage Vcoil, and back electromotive force B-EMF;

[0020]FIG. 3 is a diagram showing a phase relationship among backelectromotive force B-EMF developed in coils Lm(U), Lm(V), and Lm(W) inthe equivalent circuit of FIG. 1, coil voltage Vcoil applied across thecoils, and applied voltage Vinput by the coil drive voltage sourcesVinput(U), Vinput(V), and Vinput(W);

[0021]FIG. 4 is a diagram showing an example of drive waveforms appliedto individual phases of a three-phase brushless motor by a motor drivingcontrol circuit to which the present invention is applied;

[0022]FIG. 5 is a timing diagram showing a mutual relationship ofdriving modes of coils of individual phases of a three-phase motor and aswitching timing;

[0023]FIG. 6 is a timing diagram enlarging a range from 90 to 270degrees of FIG. 5 to show a mutual relationship of driving modes ofcoils of individual phases, a switching timing, and duty changes of SPphase;

[0024]FIG. 7 is a block diagram showing one embodiment of a drivingcontrol circuit of a three-phase brushless motor to which the presentinvention is applied;

[0025]FIG. 8 is a pattern diagram showing duty production patterns of SPphase in the motor driving control circuit of the embodiment of FIG. 7;

[0026]FIG. 9 is a flowchart showing an example of the procedure forproducing the duty of SP phase according to the patterns of FIG. 8;

[0027]FIG. 10 is a diagram for explaining changes of the duty of SPphase produced according to the procedure of FIG. 9;

[0028]FIG. 11 is a diagram showing waveforms of a PWM signal supplied toan output transistor driving a coil of SP phase produced according tothe procedure of FIG. 9;

[0029]FIG. 12 is a block diagram showing major parts of a secondembodiment of a three-phase brushless motor driving control circuit towhich the present invention is applied;

[0030]FIG. 13 is a timing diagram showing a relationship between acurrent indication value and current fluctuations developing when themotor coils are driven with sine waveforms, in the second embodiment ofthe three-phase brushless motor driving control circuit to which thepresent invention is applied;

[0031]FIG. 14 is a block diagram showing a configuration of a hard diskdrive as one example of a system employing the motor driving controlcircuit to which the present invention is applied; and

[0032]FIG. 15 is a diagram showing an example of a drive waveformapplied to coils of individual phases by a driving control circuit of aconventional three-phase brushless motor.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0033] Hereinafter, preferred embodiments of the present invention willbe described with reference to the accompanying drawings.

[0034] Before describing specific embodiments of the present invention,a driving principle of motor coils of the present invention will bedescribed using FIGS. 1 to 3. FIG. 1 shows a driving circuit in athree-phase brushless motor and an equivalent circuit of the motor. InFIG. 1, Lm(U), Lm(V), and Lm(W) respectively denote stator coils ofthree phases U, V, and W phases of a motor MT. Rm(U), Rm(V), and Rm(W)respectively denote internal resistances of phase coils Lm(U), Lm(V),and Lm(W). B-emf(U), B-emf(V), and B-emf(W) respectively denote backelectromotive forces of the phase coils Lm(U), Lm(V), and Lm(W). Ron(U),Ron(V), and Ron(W) respectively denote on resistances of outputtransistors making up a phase current output circuit that feeds currentsthrough the coils Lm(U), Lm(V), and Lm(W). Vinput (U), Vinput (V), andVinput(W) respectively denote drive voltage sources applied to thecoils.

[0035]FIG. 3 shows a phase relationship among waveforms of backelectromotive forces B-EMF developed in the coils Lm(U), Lm(V), andLm(W) in the equivalent circuit of FIG. 1, coil voltage Vcoil appliedacross the coils, and applied voltage Vinput by the coil drive voltagesources Vinput(U), Vinput(V), and Vinput(W). When AC drive currents withthe same phase as the back electromotive forces B-EMF are fed throughthe coils, the greatest torque is obtained.

[0036] However, even if drive voltages with the same phase as the backelectromotive forces B-EMF are applied to the coils, a phase lag occursin currents Icoil actually flowing through the coils because of internalresistance of the coils. Accordingly, as shown in FIG. 3, it isdesirable that coil voltage Vcoil of each phase is applied so that itsphase is Δθ coil ahead of that of the back electromotive forces B-EMFdeveloped in the coils Lm(U), Lm(V), and Lm(W), to match the phase ofcoil current Icoil to that of the back electromotive forces B-EMF. Sincevoltages Vinput applied by the drive voltage sources Vinput(U),Vinput(V), and Vinput(W) from outside the coils are also out of phasewith the coil voltages Vcoil of the individual phases, phase differencesmust be considered to decide the phases of the drive voltage sourcesVinput(U), Vinput(V), and Vinput(W).

[0037] A phase lead amount Δθ coil of the coil voltage Vcoil withrespect to the phase of the back electromotive forces B-EMF isrepresented by the following expression (1).

Δθcoil=tan⁻¹({overscore (ω)}·Lm/Ron+Rm)=tan⁻¹{(2π·fB-EMF)·Lm/(Ron+Rm)}  (1)

[0038] Δθ coil varies in value, depending on a motor used. In theexpression (1), Lm denotes coil inductance and fB-EMF denotes thefrequency of the back electromotive force B-EMF, that is, a requirednumber of revolutions of a motor.

[0039] Next, assuming that a difference between the phase of the backelectromotive forces B-EMF of the coils and the phase of the drivevoltage sources Vinput(U), Vinput(V), and Vinput(W) is Δθ, the abovedescribed applied voltage Vinput is given as a synthetic vector of thecoil voltage Vcoil and the back electromotive forces B-EMF that arerepresented by vector, as shown in FIG. 2. Hence, if inductance Lm andinternal resistance Rm of the coils are determined from the motor used,a phase difference Δθ coil can be derived from the expression (1) and Δθcan be obtained from a vector diagram of FIG. 2. Accordingly, if drivewaveforms are formed by setting the phases of the drive voltage sourcesVinput(U), Vinput(V), and Vinput(W) to be Δθ ahead of that of the backelectromotive forces B-EMF, the greatest torque can be obtained. Thephase of the back electromotive forces B-EMF developed in the coils canbe obtained by detecting a zero-cross point of the back electromotiveforces.

[0040] In a motor driving circuit of an embodiment described below, anoutput transistor is controlled so that drive voltage waveforms of thephase relationship as described above are applied to coils. Moreover,the output transistor is controlled by a PWM (pulse width modulation)system. In other words, a gate terminal of the output transistor iscontrolled by a PWM-controlled signal (pulse), whereby drive voltagewaveforms of the above described phase relationship are applied to thecoils.

[0041] As described previously, drive voltage waveforms applied to thecoils are desirably sine waveforms and their phases desirably have atiming as shown in FIG. 2. However, even if the coils are driven so asto satisfy the above condition, when the drive voltage waveforms shownin FIG. 3C are formed, if potential VCT of center tap CT, which is acommon connection terminal of coils of three phases, is kept constantand sine waveforms with the potential VCT as a center potential areformed and applied to the coils, power loss will occur in a portionhatched in FIG. 3C.

[0042] Accordingly, to reduce the power loss, we thought that apotential VCT of the center tap CT is set to be not fixed but floatingso that a coil drive voltage around a portion in which a drive waveformof each phase swings to its maximum amplitude is forcibly set to asource voltage Vcc or ground potential GND (=0V). FIG. 4A shows awaveform produced when a coil drive voltage around a portion in which adrive waveform of each phase swings to its maximum amplitude is forciblyset to the source voltage GND (=0V). FIG. 4B shows a waveform producedwhen a coil drive voltage around a portion in which a drive waveform ofeach phase swings to its maximum amplitude is forcibly set to the sourcevoltage Vcc.

[0043] It will be understood from FIG. 4 that the case (A) of FIG. 4eliminates power loss at a lower hatched portion in FIG. 2C and the caseof (B) eliminates power loss at an upper hatched portion in FIG. 2C.Accordingly, by using the drive waveforms of FIG. 4A or FIG. 4B, higherpower efficiency can be obtained than the case where a potential VCT ofthe center tap CT is fixed to drive the coils with sine waveforms asshown in FIG. 3C. In FIGS. 4A and 4B, it is because a potential VCT ofthe center tap CT floats that waveforms at portions not set to Vcc orGND appear to be not sine waveforms. Use of the potential VCT of thecenter tap CT as reference, that is, differences between-the potentialVCT of the center tap CT and the potential of individual waveforms tellthat the drive waveforms change according to sine waveforms.

[0044] In this embodiment, the above described driving system is furtheradvanced to the system of using waveforms as shown in FIG. 4C fordriving. Employing this system contributes to simplification of hardwareconfiguration. Waveforms of FIG. 4C are formed by combining waveformscut from portions from 0 to 37.5 degrees, 97.5 to 157.5 degrees, 217.5to 277.5 degrees, and 337.5 to 360 degrees from FIG. 4A and portions of37.5 to 97.5 degrees, 157.5 to 217.5 degrees, and 277.5 to 337.5 degreesfrom FIG. 4B.

[0045] Cutting is not performed in units of 60 degrees such as 0 to 60degrees, 60 to 120 degrees, 120 to 180 degrees, and 180 to 240 degrees,and so forth. This is because, as seen from FIG. 1, 60, 120, 180, 240,and 300 degrees are respectively zero-cross points of back electromotiveforce, and in this embodiment, as described later, since zero-crosspoints of back electromotive force of the coils are detected to performdriving control, switching of currents of individual phases at suchpositions causes noise to occur in back electromotive force, disablingcorrect detection of zero-cress positions.

[0046] Next, a description will be made of a specific method of formingthe drive waveforms as shown in FIG. 4. In FIG. 4C, the symbols “SP”,“PWM”, and “F” provided in the vicinity of a waveform of each phaseindicate the type of a method of forming each waveform. A differentsymbol indicates a different formation method. Hereinafter, a method offorming each waveform will be described in order.

[0047] First, a waveform marked with the symbol “F” is formed byforcibly driving an output transistor into a full amplitude level.Specifically, the output transistor driving a coil of a phasecorresponding to a waveform marked with the symbol “F” is applied with acontrol signal of high level to its gate terminal continuously for arequired time (corresponding to the length of the F waveform), therebyapplying Vcc (e.g., 12V) or GND (0V) to a driving terminal of the coil.

[0048] Next, a waveform marked with the symbol “SP” is produced byoperations in an arithmetic circuit and formed by the output transistorbeing driven by a PWM-controlled signal. As shown in FIG. 4, waveformsmarked with “SP” exist two for each of a right upward direction and aright downward direction in a range of an electrical angle 60 degreescut as described previously, and are of identical shape or verticallysymmetric shape. Therefore, they are produced by only two arithmeticexpressions. If only waveforms of a right upward direction and a rightdownward direction are formed by operations, other waveforms or part ofwaveforms can be formed by feedback control based on current detectionor full amplitude driving of the output transistor.

[0049] Thereby, in comparison with the conventional system of formingwaveforms throughout 360 degrees according to ROM data, the system ofthis embodiment forms waveforms more easily and reduces the amount ofhardware. A specific example of an operation method by an arithmeticcircuit will be described in detail later; for waveforms marked with thesymbol “SP”, PWM signals are formed which turn the output register on oroff by 16 or 32 pulses within a range of an electrical angle 60 degrees.Specifically, pulse width is controlled to become gradually wider forright upward portions and gradually narrower for right downwardportions.

[0050]FIG. 5B shows by which of “F”, “PWM”, and “SP” methods outputtransistors driving coils of three phases U, V, and W in which backelectromotive force B-EMF changes as indicated by (A) form waveforms ata proper timing. In the drawing, “upper arm” denotes a transistor of thepower voltage Vcc side of an output transistor of a corresponding phase,and “lower arm” denotes an output transistor of the GND side. A box thatis described across the boundary between “upper arm” and “lower arm” andmarked with the symbol “D” denotes that both an output transistor of theVcc side and an output transistor of the GND side are turned off. Thereason that periods are thus provided in which both an output transistorof the Vcc side and an output transistor of the GND side are turned offis to eliminate influence of drive voltages applied to coils whenzero-cross points of back electromotive forces are detected, in orderthat only the back electromotive forces are observed to detect thezero-cross points.

[0051]FIG. 6A is an enlarged view of duty changes of pulses of a signalfor driving an output transistor of PWM phase subjected to feedbackcontrol based on current detection in a portion of 90 to 270 degrees ofFIG. 5 showing waveforms in a range from 0 to 360 degrees and dutychanges of pulses of a signal for driving an output transistor of SPphase controlled by operations in an arithmetic circuit. The dutycontrol applies to not all cases and phases are automatically adjustedas shown by the arrow A according to the magnitude of output current.The phase adjustment is made based on coil current values detected inthe range of the first preceding 60 degrees. FIG. 6B is an enlarged viewof a portion from 90 to 270 degrees of FIG. 5B showing the timings ofwaveform forming methods in the range from 0 to 360 degrees.

[0052] Next, a waveform marked with “PWM” is formed based on a currentdetection and current comparison function of a motor driving controlcircuit of the embodiment. Specifically, the motor driving controlcircuit of the embodiment is provided with a current detection resistorRNF provided so that the sum of currents flowing through three coils Lu,Lv, and Lw flows to detect a total of them, and a current detectiondifferential amplifier that detects a potential difference across thecurrent detection resistor RNF to detect the magnitude of current. Tocontrol an output current, a PWM signal is produced which detects adifference between a coil current value detected by the currentdetection differential amplifier and a current indication value suppliedfrom a controller (CPU) (not shown) and drives the output transistors sothat the difference is 0.

[0053] For example, when a detected current is smaller than the currentindication value, the duty of the PWM signal is increased to allow morecurrent to flow through the coils, while, when the detected current islarger than the current indication value, the duty of the PWM signal isreduced to decrease current flowing through the coils. By repeating thisoperation, a waveform marked with the symbol “PWM” is formed. Dutycontrol of the PWM signal is performed based on the magnitude of outputcurrent detected in the preceding cycle. Thereby, the phase of dutycontrol of the PWM signal, that is, the phase of sawtooth waveform ofFIG. 6A is automatically adjusted according to an output currentdetected in the preceding cycle.

[0054] Furthermore, in this embodiment, waveforms in the range ofelectrical angle 60 degrees are formed by, e.g., 16 PWM pulses. In otherwords, the output transistors are turned on and off 16 times by 16pulses formed when a rotor rotates by an electrical angle of 60 degrees,and the respective widths of the 16 pulses are changed according to thedetected current value, whereby waveforms marked with the symbol “PWM”are formed. Since such drive pulse feedback control based on currentdetection has been performed by a motor driving control circuit of theconventional PWM control system as well, a drive waveform applied to anyone coil of three phases by the same circuits and procedure asconventional ones.

[0055]FIG. 7 shows an embodiment of the present invention effectivelyapplied to a driving control circuit of a spindle motor used in a harddisk storage apparatus. The whole circuit shown in FIG. 7 is formed onone semiconductor substrate such as a single-crystal silicon, exceptcoils Lu, Lv, and Lw of the motor.

[0056] In FIG. 7, the reference numeral 11 designates a current outputcircuit successively feeding current to the coils Lu, Lv, and Lw of athree-phase brushless motor; 12, an output current control circuit thatproduces a PWM signal for controlling an output current and supplies itto the current output circuit 11; RNF, a current detection resistorconnected to the current output circuit 11 so that the sum of currentsflowing through the three coils Lu, Lv, and Lw flows to detect a totalof them; 13, a current detection differential amplifier that detects apotential difference across the current detection resistor RNF to detectthe magnitude of current; and 14, an AD conversion circuit that performsAD conversion for an output voltage of the current detectiondifferential amplifier to produce a digital signal.

[0057] Reference numeral 15 designates a back electromotive forcedetecting circuit that detects back electromotive forces of the coilsLu, Lv, and Lw developing in output terminals u, v, and w of the currentoutput circuit 11, and center tap CT to output a signal indicating azero-cross point; 16, a phase difference detecting circuit that detectsa phase difference between a signal indicating a zero-cross point ofback electromotive force outputted from the back electromotive forcedetecting circuit 15 and a signal indicating a zero-point of an outputcurrent outputted from the output current control circuit 12; 17, a loopfilter that performs phase compensation of a main line; and 18, anoscillation circuit that oscillates at a frequency (about 100 kHz)corresponding to a value (digital code) of the loop filter 17. An outputof the oscillation circuit 18 is used as a reference clock for producingthe PWM signal in the output current control circuit 12.

[0058] PLL (phase locked loop) is formed by a feedback route establishedby the phase difference detecting circuit 16, loop filter 17,oscillation circuit 18, output current control circuit 12, and phasedifference detecting circuit 16 back from the output current controlcircuit 12. The PLL controls oscillation operation of the oscillationcircuit 18 so that the phase of a signal indicating a zero-cross pointof back electromotive force outputted from the back electromotive forcedetecting circuit 15 matches the phase of a signal outputted from theoutput current control circuit 12, thereby locking the frequencies ofvoltage waveforms (1 to 2 kHz) applied to the coils.

[0059] Reference numeral 19 designates an AD conversion circuit thatperforms AD conversion for a back electromotive force outputted from theback electromotive force detecting circuit; 20, a conduction startcontrol circuit that decides a conduction start phase, based on a backelectromotive force induced in a nonconduction phase and detected by theback electromotive force detecting circuit 15 when a short pulse towhich the rotor does not respond is fed from one phase to another by thecurrent output circuit 11, based on an output of the AD conversioncircuit 19 when the motor is standing; 21, a serial port that sends andreceives data to and from a microcomputer (CPU) (not shown).

[0060] The serial port 21 receives a serial clock SCLK supplied from theCPU, a current indication value of a spindle motor, and informationabout an operation mode, and produces control signals inside the drivingcontrol circuit, based on received mode information.

[0061] Reference numeral 22 designates a sequencer that controls thewhole of circuits shown in FIG. 7; 23, an arithmetic circuit thatproduces a duty control signal for forming drive waveforms of SP phase;24, a current difference detecting circuit that detects a differencebetween a coil current value detected by the current detectiondifferential amplifier 13 and a current indication value supplied viathe serial port 21 from the CPU; and 25, a filter that produces a valuecorresponding to a current difference detected based on an output of thecurrent difference detecting circuit 24 while making phase compensation.Current difference information outputted from the filter 25 and waveforminformation produced in the arithmetic circuit 23 are supplied to theoutput current control circuit 12, where a PWM signal is produced todrive the output transistors and supplied to the current output circuit11 to control output currents to be fed to the coils.

[0062] Output current Iout is represented by

Iout={(Vcc×Duty)−Bemf}/RL

[0063] where Duty is the duty (ratio of pulse width to one cycle) of PWMsignal, Bemf is coil back electromotive force, and RL is coilresistance. Accordingly, changes of PWM signal cause coil output currentIout to be controlled according to the above expression.

[0064] Next, a more specific method of producing waveforms (hereinafterreferred to as waveforms of SP phase) marked with the symbol SP in FIG.4C will be described using FIG. 8.

[0065] First, a description is made of the case where coil backelectromotive force B-EMF and coil voltage Vcoil are not out of phasewith each other. Suppose that an output current and a current indicationvalue from the CPU match and the duty of a control signal for producingwaveforms (hereinafter referred to as waveforms of PWM phase) markedwith the symbol PWM in FIG. 4C is constant (e.g., 70%). FIGS. 8A and 8Bshow a relationship between back electromotive force B-EMF in that caseand the duty of a control signal for producing waveforms of SP phase.FIG. 8A shows the duty of PWM phase at the left scale and the duty of SPphase in a direction opposite to PWM phase at the right scale,representing changes in PWM phase duty (constant) and SP phase duty,correspondingly to the respective scales.

[0066] In FIG. 8, in an electrical angle of 90 degrees, the backelectromotive force of U phase is zero, just in the middle of the backelectromotive forces of V phase and W phase. At this time, V phase isPWM phase, U phase is SP phase, and V phase is a phase (hereinafterreferred to as F phase) driven into full amplitude. Accordingly, theduty of U phase, which is SP phase, is 65% of a complement D1(=100−D0/2) for 100% of just the half (D0/2) of the duty (D0 (=70%) of Vphase, which is PWM phase. Since waveforms of U phase change to fullamplitude in a section from 65% to 100%, the duty may be changed from65% to 100%. In this embodiment, since duty changes at this time couldbe linearly made with only small errors, linear changes arealternatively employed to simplify control.

[0067] If a waveform of U phase reaches duty 100% at 120 degrees,thereafter, U phase is switched to the F phase of full amplitudedriving, W phase, which has been hitherto F phase, is switched to SPphase, and the duty of the control signal is linearly changed from 100%to 65%. Phase switching is made again at an electrical angle of 150degrees such that V phase, which has been hitherto PWM phase, isswitched to SP phase, the duty of the control signal is linearly changedfrom 65% to 100%, W phase, which has been SP phase, is switched to Fphase, and U phase, which has been F phase, is switched to PWM phase;this is continued up to an electrical angel of 180 degrees. At anelectrical angle of 180 degrees, F phase, which has been hitherto Fphase, is switched to SP phase, the duty of the control signal islinearly changed from 100% to 65%, and V phase, which has been SP phase,is switched to F phase. At this time, U phase is left to be PWM phase.

[0068] The above waveforms are true for the case where coil backelectromotive force B-EMF and coil voltage Vcoil are not out of phasewith each other. If coil back electromotive force B-EMF and coil voltageVcoil are out of phase with each other, the waveforms are formed asshown in FIG. 8C. That is, with the same tilt as the tilt of duty changeof each SP phase in FIG. 8B, a starting point is advanced by phase Δθ tocontrol the duty of each SP phase. By this arrangement, the phase of acoil voltage Vcoil of each phase leads the phase of back electromotiveforce B-EMF and coil current Icoil is driven in phase with backelectromotive force B-EMF, so that the greatest torque can be produced.

[0069] In the case of FIG. 8B, phase switching takes place at electricalangles of 30, 90, 150, 210, 270, and 330 degrees, and these pointscorrespond to zero-cross points of back electromotive force B-EMF. Forthis reason, phase switching at these points may cause noise to occur inback electromotive force and disable correct detection of zero-crosspoints. Accordingly, as shown in FIG. 8D, it is desirable to controlduties so as to delay phase switching timing by Δoffset (e.g., anelectrical angle 7.5 degrees).

[0070] Since a V-shaped waveform of FIG. 8D highly resembles a waveformof SP phase in a range from 100 to 160 degrees in FIG. 4C, it isunderstood that a waveform similar to a desired waveform (sine waveformas viewed from the center tap) can be produced by the above describedduty control method. It can be easily determined that waveforms of SPphase in other portions, which are vertically symmetrical, can berealized by reversing the positive/negative relationship of the abovedescribed duty control. In the motor driving control circuit of theembodiment of FIG. 7, the above described duty control is achieved bythe arithmetic circuit 23 and a PWM control circuit within the outputcurrent control circuit 12 in coordination.

[0071] Next, the procedure of operations in the arithmetic circuit 23for producing waveforms of the above SP phase is described using aflowchart of FIG. 9. The meanings of variables used in the procedure bythe flowchart are shown in FIG. 10. As seen from FIG. 10, PWM controlfollowing the flowchart is not continuous but is performed in 16 stagesaccording to 16 PWM pulses in one conduction period (electrical angle of60 degrees). The number of PWM pulses in one conduction period isarbitrary.

[0072] The PWM control circuit, which produces a predetermined number(e.g., 16) of PWM pulses in a conduction period of each phase andapplies them to an output transistor, successively adds on time (e.g.,high level period) of the 16 PWM pulses in one conduction period to findtotal on time Ton-total, and calculates an average value PWMave bydividing the total time (Ton-total) by the number of pulses DIV at phaseswitching (step S1). An output value of the AD conversion circuit 14 inone conduction period is successively added and the total of them isdivided by the number of pulses DIV at phase switching to find anaverage value Itotalave of a total output current (step S2).

[0073] Next, a coefficient CIADJ (=Δθ/Itotal) inputted from the CPU viaa serial port and the average output current Itotalave calculated instep S2 are multiplied to obtain a phase lead amount Δθ1 of an appliedvoltage Vinput applied to a coil (step S3). Δθ and Itotal, instead ofthe coefficient CIADJ, may be given from the CPU to obtain a coefficientby operations in the motor driving control circuit.

[0074] In the next step S4, a value Δθ2 (=Δθ1−Δoffset) is calculated bysubtracting a delay amount Δoffset of phase switching timing fromzero-cross point from the phase lead amount Δθ1 obtained in step S3. Theaverage value PWMave of total on time of PWM pulses calculated in stepS1 is divided by the number of pulses DIV to obtain an average dutychange amount Δndown (=PWMave/DIV) per PWM pulse. In the next step S6,the average duty change amount Δndown obtained in step S5 is multipliedby Δθ2 obtained in step S4 to find an decrease amount ΔCNT from theaverage value PWMave of total on time of PWM pulses.

[0075] The average value PWMave of total on time of PWM pulses is halvedto obtain a loopback point duty (D1 of FIG. 8B) of the SP phase in thecase where it is assumed that there is no phase lag, and the value issubtracted by the decrease amount ΔCNT obtained in step S6 to calculatethe duty SSN0 of a PWM pulse applied to a first SP phase after phaseswitching (step S7). Thereafter, the duty of PWM pulse a second time orlater, namely, on time SSNd is decided by subtracting the change amountΔndown obtained in step S5 from on time SSNd-1 of a previous PWM pulse(step S8).

[0076] In the next step S9, it is judged whether on time SSNd decided instep S8 is equal to or smaller than 0, and step S8 is repeated untilSSNd is equal to or smaller than 0, whereby the duties of SP phase in adown period indicated by the symbol Tdown in FIG. 10 are successivelyoutputted. If on time SSNd is equal to or smaller than 0, control istransferred to step S10, where a predetermined change amount Δndown1 isadded to on time SSNu-1 of the previous PWM pulse to decide the duty ofthe next PWM pulse, namely, on time SSNu.

[0077] In the next step S11, it is decided whether the number N ofproduced pulses reaches the number DIV of pulses in one conductionperiod, and step S10 is repeated until N and DIV match, whereby theduties of SP phase in an up period indicated by the symbol Tup in FIG.10 are successively outputted. A line indicated by a dashed line A is aduty change line of the SP phase in the case where it is assumed thatthere is no phase lag, and corresponds to the waveform in FIG. 8B. Theduties SSNd and SSNu calculated in the steps S8 and S10 are alloutputted as one's complement numbers (1-SSNd) or (1-SSNu). This is doneto convert duties calculated at the left scale of FIG. 8 to the rightscale.

[0078]FIG. 11A shows PWM pulses in the down period Tdown of the SP phaseof FIG. 10 produced based on the duty SSNd calculated in step S8 of FIG.9, and FIG. 11B shows PWM pulses in the up period Tup of the SP phase ofFIG. 10.

[0079] The PWM pulses of FIG. 11A are applied to the gate terminal of anoutput transistor (N-MOS) of the Vcc side driving the coil (Lv) formingSP phase in a period from 37.5 to 55 degrees of FIG. 4C. If thetransistor is P-MOS, inversion signals of the PWM pulses of FIG. 11A areapplied to the gate terminal. The PWM pulses of FIG. 11A are applied tothe gate terminal of an output transistor (NMOS) of the GND side drivingthe coil (Lu) forming SP phase in a period from 97.5 to 115 degrees ofFIG. 4C.

[0080] The PWM pulses of FIG. 11B are applied to the gate terminal of anoutput transistor (N-MOS) of the Vcc side driving the coil (Lu) formingSP phase in a period from 55 to 97.5 degrees of FIG. 4C. If thetransistor is P-NOS, inversion signals of the PWM pulses of FIG. 11B areapplied to the gate terminal. The PWM pulses of FIG. 11B are applied tothe gate terminal of an output transistor (N-MOS) of the GND sidedriving the coil (Lw) forming SP phase in a period from 115 to 157.5degrees of FIG. 4C. By the above described method, waveforms of SP phasethat are vertically symmetrical can be formed according to an identicalprocedure, using identical values.

[0081]FIG. 12 shows a configuration of major parts of a motor drivingcontrol circuit in a second embodiment of the present invention.

[0082] As described previously, the motor driving control circuit of thefirst embodiment is provided with the current detection resistor RNF fordetecting a total current flowing through the three coils Lu, Lv, and Lwand the differential amplifier 13, wherein a difference between adetected coil current value and a current indication value supplied fromthe controller (CPU) outside the drawing is detected, and a PWM signalis produced to drive the output transistor so as to make the differencezero so that output current fed through the coils is subjected tofeedback control. On the other hand, in the motor driving controlcircuit of the present embodiment, since coils of three phases of themotor are driven with three sine waveforms that are 120 degrees out ofphase with one another, a total current Itotal flowing through the motorfluctuates and forms a rippled waveform indicated by a solid line B inFIG. 13.

[0083] If the total current is detected by the current detectionresistor RNF and the differential amplifier 13 and compared with acurrent indication value SPNCRNT (constant within a short time) givenfrom the CPU, judging that an error occurs, the feedback control systemof the output current control circuit 12 reacts to the ripple andchanges output current. Since there is a delay in the current controlsystem, torque ripple becomes worse.

[0084] Accordingly, in the embodiment of FIG. 12, an error currentdetecting circuit 24 is provided with a correction arithmetic circuit 26that corrects a current indication value by multiplying a currentindication value SPNCRNT given from the CPU by a coefficient, and aselector 27. The coefficient multiplied by the current indication valueSPNCRNT is a value such as, e.g., 1.1, according to average regulationof output current. The selector 27 selects between a value with acurrent indication value SPNCRNT multiplied by a coefficient and a valuewith the current indication value SPNCRNT not multiplied by acoefficient and supplies the value to an add circuit 28 that finds adifference between the value and an output of the AD conversion circuit.

[0085] Switching timing of the selector 27 can be automatically obtainedfrom phase switching timing of the output current control circuit 12.Specifically, taking delay in the control system into account, theselector 27 may be subjected to switching control so that the selector27 selects a value with a current indication value SPNCRNT multiplied bya coefficient as in FIG. 13A in accordance with the timing when the ADconversion circuit 14 outputs current values corresponding to ridgedportions of the total current Itotal indicated in FIG. 13B, and selectsa value with the current indication value SPNCRNT not multiplied by acoefficient in accordance with the timing when the AD conversion circuit14 outputs current values corresponding to valley portions of the totalcurrent Itotal. The coefficient multiplied by the current indicationvalue SPNCRNT may be a value smaller than 1 such as e.g., 0.9, to switchthe selector in the reverse timing of the above.

[0086] As in this embodiment, by changing the current indication valueSPNCRNT according to the fluctuation of coil total current Itotal,reaction of the control system to ripples of coil current can beweakened, with the result that torque ripples resulting from driving themotor with a sine waveform can be reduced. Although, in this embodiment,a current indication value SPNCRNT is changed at two levels, pluralcorrection arithmetic circuits 26 that corrects a current indicationvalue by multiplying a current indication value SPNCRNT by a coefficientmay be provided and appropriately selected by the selector 27 accordingto the fluctuation of total current Itotal so that the currentindication value SPNCRNT is changed at three levels or more.

[0087]FIG. 14 is a block diagram showing a configuration of a hard diskdrive as one example of a magnetic disk system including a spindle motorcontrol system employing a motor driving control circuit to which thepresent invention is applied, and a magnetic head driving controlsystem.

[0088] In FIG. 14, 210 designates a spindle motor driving controlcircuit, which is configured as shown in FIG. 7, drives and controls thespindle motor 310, and rotates magnetic disk at a predetermined speed.The spindle motor driving control circuit 210 operates according tocontrol signals such as a current indication value SPNCRNT supplied froma controller 260 comprising a microcomputer and performs servo controlfor the spindle motor 310 so as to keep relative speed of a magnetichead constant.

[0089] Reference numeral 320 designates an arm having a magnetic head(including a write magnetic head and a read magnetic head) HD and 330designates a carriage rotatably holding the arm 320. The voice coilmotor 340 moves the carriage 330 to move the magnetic head, and a VCMdriving circuit 100 performs servo control for the voice coil motor 340to align the center of the magnetic head with the center of track.

[0090] Reference numeral 220 designates a read/write IC that amplifiescurrent corresponding to a magnetic change to send a read signal to asignal processing circuit (data channel processor) 230 or amplifies awrite pulse signal from the signal processing circuit 230 to outputdrive current of the magnetic head HD. Reference numeral 240 designatesa hard disk controller that gets read date sent from the signalprocessing circuit 230 to perform error correcting processing, andperforms error-correcting encoding processing for write data from a hostto output the result to the signal processing circuit 230. The abovedescribed signal processing circuit 230 performs modulation/demodulationprocessing suitable for digital magnetic recording and signal processingincluding waveform shaping with magnetic recording characteristics inmind, and reads position information from a read signal of the magnetichead HD.

[0091] Reference numeral 250 designates an interface controller thatperforms data exchange and control between this system and externalapparatuses, and the hard disk controller 240 is connected to a hostcomputer such as a microcomputer of a personal computer body via theinterface controller 250. Reference numeral 270 designates a cachememory for temporarily storing read data read at high speed frommagnetic disk. A system controller 260 comprising a microcomputer judgesan operation mode from a signal supplied from the hard disk controller240, controls various parts of the system according to the operationmode, and calculates a sector position and the like from addressinformation supplied from the hard disk controller 240.

[0092] As described above, the present invention made by the inventorhas been described in detail based on preferred embodiments. It goeswithout saying that the present invention is not limited to the abovedescribed preferred embodiments, but may be modified in various wayswithout departing from the spirit and the scope of the presentinvention. For example, in the motor driving circuit of the abovedescribed embodiments, although the sensorless method is employed todetect a rotor stop position and decide a conduction start phase bydetecting back electromotive force, a rotor stop position may bedetected using a hole sensor or the like. The motor may be not athree-phase motor but multiple-phase motor.

[0093] Although, in the embodiments, waveforms of SP phase are producedby operations in an arithmetic circuit, a memory to store datacorresponding to waveforms may be provided so that waveforms areproduced by successively reading the data from the memory. Moreover,although, in the embodiments, a MOS transistor is used as an outputtransistor, a bipolar transistor can be used as an output transistor.Moreover, although, in the embodiments, the full-wave driving method isdescribed, the present invention can apply to the half-wave drivingmethod also.

[0094] Although the present invention has been described as toapplication to a motor driver apparatus of a hard disk storageapparatus, which is an application field of the present invention, thepresent invention is not limited to such a field and can be widely usedin a motor driving control apparatus driving brushless motors such as,e.g., a motor for rotating a polygon mirror of a laser beam printer andan axial fan motor.

[0095] Effects obtained by typical ones of inventions disclosed by thepresent patent application are briefly described below.

[0096] According to the present invention, currents of sine waveformscan be fed through coils by a relatively small-sized circuit. With thisconstruction, highly dense magnetic storage can be realized with lessrotation variations and a magnetic disk unit provided with a spindlemotor rotating at a low noise level can be achieved.

What is claimed is:
 1. A magnetic disk storage apparatus comprising: afirst motor for rotating magnetic disk; a magnetic head for readinginformation from recording tracks on the magnetic disk; and a firstmotor driving control circuit for controlling drive currents of thefirst motor, wherein the first motor is a multi-phase brushless motor inwhich the potential of a center tap of the multi-phase brushless motoris made to be floating, and wherein a driving control circuit of thefirst motor performs driving by feedback control so that a coil of oneof the phases is driven with an amplitude at which an applied voltagebecomes equal to a source voltage, a coil of a second phase is drivenwith gradually changing voltages so that a current of sine waveform isdelivered; and a coil of a third phase is controlled so that a totalcurrent flowing through all coils becomes a predetermined current value.2. The magnetic disk storage apparatus according to claim 1, wherein thefirst motor driving control circuit includes an arithmetic circuit thatproduces by predetermined operations a signal driven with graduallychanging voltages so that a current of sine waveform is delivered. 3.The magnetic disk storage apparatus according to claim 2, wherein thefirst motor driving control circuit produces as a PWM signal a signaldriven with gradually changing voltages so that a current of sinewaveform is delivered.
 4. The magnetic disk storage apparatus accordingto claim 3, wherein the first motor driving control circuit produces asa PWM signal a signal driven with the feedback control.
 5. The magneticdisk storage apparatus according to claim 1, wherein coil currents fedthrough coils of individual phases by the first motor driving controlcircuit are formed to have phases that are an predetermined electricalangle corresponding to coil inductance and internal resistance ahead ofthe phases of back electromotive forces induced in the coils.
 6. Themagnetic disk storage apparatus according to claim 5, wherein the firstmotor driving control circuit drives coils of individual phases so thatphase switching timing is off zero-cross points of the backelectromotive forces.
 7. The magnetic disk storage apparatus accordingto claim 1, wherein the first motor driving control circuit producessignals driven with gradually changing voltages by identical operationseven if phases driven by the signals are different from each other sothat currents of sine waveforms are delivered.
 8. The magnetic diskstorage apparatus according to claim 1, including the first motordriving control circuit and a controller controlling the first motordriving control circuit, wherein the first motor driving control circuitis constructed to perform control so that the total of currents fedthrough the coils of the phases matches a current indication valuesupplied from the controller, and includes a current indication valuecorrecting circuit that corrects the current indication value, takinginto account fluctuations of the total current produced by the currentsfed through the coils of the phases being changed according to sinewaveforms.
 9. The magnetic disk storage apparatus according to claim 1,wherein the first motor is a three-phase brushless motor.